Negative feedback detector



J. W. MCRAE NEGATIVE FEEDBACK DETECTORS March 3o, 1943.

Filed Feb. 11, 1942 3 Sheets-Sheet 1 3% FIG.l 3

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/NVENTOR J. W MC RAE ATTORNEY March-30,1943. J. w. MORA: Y2,315,442

NEGATIVE FEEDBACK DETEcTons ATrQRA/Ey March 30, 1943. J. w, McRAE2,315,442

NEGATIVE FEEDBACK DETECTORS Filed Feb. 11. 1942 s sheets-sheet s SIGNALOUTPUT SOURCE 0F DOUBLE B. NODULTED MODULATED CARRIER SIGNAL OUTPUT/NVENTQR J W MCR/1E Arrow/Ey l Patented Mar. 30, 1943 UNITED STATES Paraorrics l Telephone Laboratories,

Incorporated, New

York, N. Y., a corporation of New York Application February 11, 1942,Serial No. 430,383

(Cl. Z50-27) 8 Claims.

This invention relates to modulation, especially detection. Y

An object of the invention is to increase fidelity of detection ordemodulation.

In one specific aspect of the invention an amplifier operating Vatenvelope frequencies and having a large amountof negative feedback is soassociated with a self-biasing detector that the feedback not onlyreduces distortion and phase shift between the detectorv bias voltageand the amplifier output but so modifies the apparent de-r tector loadimpedance as to increase fidelity of detection. Y

Other objectsand aspects of the invention will be apparent'from thefollowing description and claims. A

Fig.` 1 is a detector circuit diagram for facilitating explanation ofthe invention; y

Fig. 1A is the envelopeA frequency equivalent circuit corresponding toFigfl;

Fig. 2 shows a detector circuit embodying a form of the invention; Y

FigfSshows a modified formV of circuit with means for feedingbackenvelope frequency currents to the exclusion of directl current;

Fig. 4 shows a generalized form of detector circuit from which circuitssuch as those of Figs. 2 and 3 canbe derived;

Figs. 5, 5A and 5B show three modified forms of detector circuitsemploying detector tubes having in addition to the anode and cathode atleast one control element; Fig. 5C shows a modification of a portion ofthe circuit of Fig. 5B;

i Fig. 6 showsa circuit adapted for modulation of a carrier waveorjdemodulation of a modulated wave; and Y Y Fig. 7 shows an envelopefrequency feedback system with the feedback path including a detectorcircuit generally similar to the detector circuit of Fig. 3.

A fundamental circuit for a self-biasing diode detector is shown in Fig.l, wherein I is a tuned radio frequency input circuit or circuit towhich the signal modulated carrier waves to be demodulated are applied,2 is a diode having inherent plate-cathode capacity Cd, R is a loadresistance across which the detected signal voltage is desired, and Ccis a blocking condenser for the detected signal but a lowv or negligibleimpedance for the input waves. The signal to distortionl ratioVv of suchdetectors is limited in two ways.

Firs t there is the non-linearity inherent in theV diode characteristicand, second, there is the effect of the frequency characteristic of theload 4impedance Z, which is shown in its simplest form in Fig. 1Awherein E is the diode bias voltage and C=Cc-l-Cd. In order to reducethe effect of diode non-linearity it is desirable to increase theabsolute value of Z. However, because of the presence of capacities Ceand Cs, such an increase in absolute value of Z tends to result in adecrease in what may be called the critical frequency of Z, i. e. thefrequency at which Z has its shunt reactance equal to its shuntresistance. Where deep modulation must be handled at relatively high`envelope frequencies, excessive reduction of this critical frequencytends to result in increased distortion. For thel detection of broadcastsignals, where the modulation at high audio frequencies is very small,no difficulty is ordinarily encountered in securing sufficiently highvalues of Z. In other applications it may be found that Z cannot be madesufficiently high to overcome diode non-linearity without making itscritical frequency too low. An example of this kind of application .isthe beta-circuit demodulator V(i. e. the demodulator in the envelopefeedback circuit) in multi-plex radio transmitter circuits employingenvelope feedback, (for instance circuits of the general type of thecircuit of Fig. 7 described hereinafter). Additional examples may befound in some forms of. measuring equipment and also possibly intelevision receivers.

vThis difficulty can be overcome in many cases with the help of anegative feedbackamplier responding tofrequencies in the envelope of theincoming signal, as for example, amplifier A in Fig. 2, wherein theimpedance Z1 across the grid and cathode of the amplifier comprisesresistance R and radio frequency by-pass capacity C'. In discussingoperation of this circuit, only the direct and envelope frequencycomponents of the diode current, and the voltages set up by thesecurrents, need be considered. The diode current id (constituted by thesecurrent components) owing through the impedance Z1 across the grid andcathode of the amplifier A, sets up across Z1 a voltage E1=z`dZ1 andthis voltage sets upa voltage Eo across the output terminals of theamplifier, tending to bias the diode to cutoff'. The resultant diodebiasvoltage vis E=E1+Eo. Designating the amplification of the amplifier,

andthe apparent diode load impedance, which is defined as the ratio ofthe diode bias voltage to diode current (averaged over a number of radiofrequency cycles), is

Z=f=z1 f1+o 2 If A is large the apparent diode load impedance istherefore approximately equal to the impedance Z1 multiplied by theamplification of the amplifier and the output voltage Ec is almostexactly in phase with and slightly less than the diode bias voltage. If,further, A is real and constant over the required band of envelopefrequencies, Z will have the same frequency characteristics as Z1, andsince Z1 may be made much smaller than Z, it may be made to have muchmore desirable characteristics than a physical impedance of themagnitude of Z. By taking ad- Vantage of this fact, distortion in thediode may be kept small, even for signals containing a wide band ofenvelope frequencies.

The negative bias on the diode, tending to cut it off, is obtained fromthe output of the amplifier rather than directly from the fiow ofcurrent through a physical impedance of magnitude Z. Thus the diodecurrent, which is only required to control the input voltage of theamplifier, may be considerably reduced from the same diode bias voltage,and since the apparent diode load impedance is the ratio of diode biasto diode current this reduction is equivalent to an increase in theapparent diode load impedance Z. More important, the fact that thisincrease in impedance is obtained by the action of a feedback amplifiermeans that the variation of Z with frequency may be made moresatisfactory than in the case of a purely passive physical impedance. Inthis way, distortion generated in the diode in many cases may be madeconsiderably smaller than in the simple diode circuits (i. e., the diodecircuits without the feedback amplifier). The circuit configurationresults in a large amount of feedback around the amplifier. In additionto its effect upon the apparent diode load impedance, this feedbacktends to reduce distortion generated in the amplifier itself. Thus, arelatively large detected output, with relativelyvlow distortion, may beobtained with only a small input. Besides the resultant high detectorinput impedance, this means that sufciently low distortion may often beobtained with smaller diodes than Would normally be required. Atultra-high. carrier frequencies this is sometimes an importantadvantage.

Because of the presence of feedback, the envelope frequency circuitsshould be designed to pass a frequency band somewhat wider than therequired envelope band, in order to facilitate proper shaping of theInu-beta loop transmission characteristic as disclosed in II. W. BodePatent 2,123,178, July 12, 1938. Therefore, to avoid difficulties inkeeping radio frequencies out of the amplifier, it is desirable to havethe highest envelope frequency low compared to the carrier frequency.

The amplifier may have any desired number of stages', provided the phaseof the feedback is maintained correct for the desired negative feedbackaction. In the envelope frequency band the phase-shift through theamplifier is 180 degrees. Although the amplifier may be capable ofresponding to direct as well as alternating currents and voltages, stillmore desirable operation, (in particular, even more desirable impedancecharacteristics), may be obtained by a modified circuit that makes useof an amplifier which does not transmit direct current but is responsiveonly to alternating voltage, as for example, the circuit of Fig. 3.

In this circuit of Fig. 3, condensers CB are blocking condensers fordirect current. A radiofrequency choke 3 isolates the plate of the diode2 from the envelope-frequency amplifier A at radio frequencies, toprevent overloading the amplifier. Re is a grid leak resistor, Rp aplate resistor, R2 a coupling resistor and RL, a load resistor for theamplifier. There is a phase reversal in transmission at envelopefrequencies through the amplifier. C1 is a radio-frequency by-passcondenser. The average diode bias corresponding to the carrier is setup'across the resistors Ri and R2 in series, the amplifier beingisolated from this voltage by the blocking condensers CB. On the otherhand, the apparent impedance presented to the alternating components ofenvelope frequency in the diode current is still given by Equation 2.The effect of the blocking condensers must of course be considered incomputing the amplification A of the amplifier A and, except at thelowest en velope frequencies, Zi now includes the effect of the gridresistor Rc at the grid of the amplifier tube, (or at the grid of thefirst amplifier tube Where the amplifier is a multistage amplifier).

This arrangement makes it possible to select the direct and alternatingcomponents of the apparent diode load impedance independently. In thisway the occurrence of the most serious type of distortion in the diodecan be avoided. This type of distortion is often known as nontrackingdistortion since it occurs when the envelope of the signal is decreasingat too great a rate for the bias voltage to follow or track When thediode load impedance consists of a simple resistance-capacitanceparallel combination, the criterion for the occurrence of this kind ofdistortion may be simply related to the time constant of the loadimpedance. However, in more complicated cases, such as those underconsideration here, another Vapproach seems desirable.

Non-tracking distortion occurs when the al ternating component of diodecurrent becomes greater than the direct, or carrier component. In thatcase, if there were to be no distortion, the diode current would have tobecome negative during a portion of the envelope cycle. Since this isimpossible, the result follows that the negative peak of the diodealternating current is cut off and during this interval the load isisolated from the incoming signal. Thus the output voltage no longerfollows the envelope but simply decreases in transient fashion.

To avoid this kind of distortion, the peak magnitude of the ratiobetween envelope frequency components and the detected carrier componentof diode current must be less than unity. For single tone modulationthis criterion may be written where 7c is the modulation factor for theincoming signal voltage, R is the resistance presented to the direct (i.e., carrier) component of diode current, and Z is the apparent diodeload impedance at the frequency of the envelope of the signal. Now Zusually falls off with increasing frequency. VThus a usually suflicientVcondition for the avoidance of non-tracking distortion, even for complexmodulation, is obtained by setting up the above criterion, using themaximum total modulation factor and the maximum frequency in theenvelope band. In many cases, this condition is much too severe and maybe relaxed considerably. However it may be taken as the basis for thefollowing discussion of the design procedure.

The first step in this procedure is to set the value for R (equal toRi-l-Rz in Fig. 3) sufficiently high to give the required low distortiondue to diode non-linearities. Then the impedance Z1 is decided upon, thecapacitance being large enough to give the required radio frequencyby-passing and the resistance then being adjusted to give a reasonablyat characteristic over the required band of envelope frequencies. Theamplifier gain necessary to raise the apparent diode load impedance to avalue approximating Rrnay then be found. Since the amplification will,in general, not be flat over the entire band, it will -be necessary touse Relation 2 in order to find the characteristics of Z. At the highestfrequency in the band, Criterion 3 will give a minimum permissible valueof Z, in terms of R and the characteristics of the expected signal. .Theaim of the design should be to make Z exceed this minimum as little asposible, consistent with necessary allowances for loss of gain in theamplifier as the tubes age. In addition, Z should be as fiat as posible,so as not to exceed the value of R greatly at the low vfrequency end ofthe required band of envelope frequencies. Such excessive valuesv of Zwould introduce a type of distortion not as yet mentioned. However,experience indicates that even with ratios of Z to R as great as 3 thistype of distortion is not important. Y

If more than a single stage is required in the amplifier, precautionsmust be taken to avoid singing around the feedback loop. The usualdesign methods may of course be employed for this purpose, and it willusually be found satisfactory to consider the diode as -a zero impedancegenerator in making such calculations.

Means should also be provided to keep excessive radio frequency voltagesout of the amplifier, for

'example filtering circuits such as shown in Fig. 7

described hereinafter. This is particularly true if the amplifier tubesare operating nearly to capacity under the given envelope frequencyconditions. Then a relatively Vsmall radio frequency voltage in theamplifier may considerably increase the distortion. The effect of thenecessary filter.- ing .circuits on the amplifier vfeedback loop must,of course, be considered in the stability calculations.

v 'A specific design example' of a detector of the type shown in Fig. 3was one for use in a l2-channel multiplex radio telephone transmitteremploying envelope-l feedback and operating on a carrier frequency of141 megacycles. A single 6L6 tube was used as the amplifier. Thisdetector was to be used (in the fashion indicated in Fig. 7 describedhereinafter) to obtain a sample of the output for feeding back throughthe beta circuit to the input of the signal amplifier, and therefore itwas important that it should introduce negligible distortion throughoutthe envelope frequency band and that its overall phase shift should benegligibly small up to frequencies of the order of a mega-cycle. Thedesign was made to permit operation over an envelope frequency band froml2 to 108 kilocycles. v

The value of R was set at 550,000 ohms, and values computed for curves(not shown) of A,

A A l Z1, and Z. At 108 kilocycles, the absolute value of Z was about 4decibels higher than R (that is, about 870,000 ohms). There wasconsequently a large safety factor against non-tracking, even for percent modulation at 108 kilocycles. In fact, this safety factor wasexcessive in this application and could have been reduced by increasingR to 870,000 ohms or more. At the lower frequencies, Z increased to amaximum value of about 1.5 megohms. On the other hand, at very highfrequencies, Z fell on" rather rapidly, the slope being nearly 12decibels per octave over a portion of the range. In this region the realpart of Z was negative. Nevertheless, because of the low internalimpedance of the diode, the circuit was found to be stable. In addition,despite the somewhat low value of R, the distortion was found to besufficiently low.

Fig. 4 shows a general form of detector circuit of which the circuits ofFigs. 2 and 3 can be considered specic types. In Fig. 4 the amplifier isshown as having a plurality of sections A1 and A2 in tandem, (each ofany desired number of stages), with a pair of output terminals 4 betweentwo of the sections. An attenuating or transmission control network cmay be provided between the amplifier output terminals 5 and the diode.Equalization (or control of the attenuation versus frequency andphaseshift versus frequency characteristics of the feedback path) toprevent singing may be provided, at least partially, in this attenuatingnetwork, and this network may include condensers (indicated in dottedlines) preventing feedback of direct current. The presence of thisnetwork provides a choice as to the outputfi. load) connection for thesyst-em. For example, the output of the system may be taken off at thediode (from terminals S, across resistor R2', or at the amplifier output(terminals 5), or at some intermediate point on the attenuating networkor at terminals For each of these conditions analysis of the operationof the system would take the same general forni as that presented inconnection with the preceding figures of the drawings.

Figs. 5, 5A and 5B show examples of ways in which multielement tubes,that is, tubes having in addition to an anode and cathode at least onedischarge control element, may be used as the detector elementscooperating with the amplifier devices. As in the case of the amplierelements of Figs. 2 to 4, the amplifiers A of Figs. 5, 5A and 5B may belinear devices, the detector elements being the only necessarilynon-linear elements of the systems. The multielement detector tubes areshown at 2 and may be, for example, screen grid tubes or suppressor gridpentodes. The ampliiiers A may be assisted to some extent in theamplication function by the gain (to envelope frequency voltages)existing in the detector tube. However since this tube is operatedaround its cut-off, its gain will probably be very small or may even benegative. In addition the cut-off in multielement tubes ordinarily isnot as sharp as in a diode, and because of this greater nonlinearity itmay be necessary to present a higher equivalent load impedance to thedetector than in the case of a diode.

'Ihe operation of these circuits is similar in general to those alreadydescribed. In each case,

the principal function of the feedback at envelope frequencies is tofacilitate the maintenance of a high load impedance for the detector andto reduce the possibility of non-tracking distortion. In addition, muchgreater output power may be obtained from the amplifier-detectorcombination for a given amount of distortion than could be obtained froma simple detector. Finally, the feedback tends to reduce distortion inthe amplier in addition to its effect on the apparent detector loadimpedance.

One feature of these circuits which is not characteristic of thecircuits using diodes is the existence of what may be an appreciableZero signal current in the detector tube. This component. of currentcould -be set up in the diode of any of the preceding circuits by theapplication of a positive direct current bias to the diode anode. Such abias is known to reduce the possibility of non-tracking in a simplediode detector but it has the disadvantage of reducing the diode inputimpedance for small signals. The circuits of Figs. 5. 5A and 5B do nothave this disadvantage and the input impedance presented to the signalwill be considerably higher than that for any type of detector using adiode. This is a result of the use of a multielement tube as thedetector, and of the fact that the detector may be operated with no gridcurrent. At the same time, as will be shown later, the no-signal currentin the platecathode circuit of the detector tube is similar to apositive bias on a diode in reducing the probability of non-trackingdistortion.

Consider first the circuit of Fig. 5, which is drawn for the case inwhich amplifier A is a direct-current amplier. With no signal, anequilibrium condition is established between the plate-cathode currentin the detector tube 2' and the direct current bias voltage developedacross cathode impedance Ze, which may be, for example, a resistance 'iand a carrier or radio frequency by-pass condenser 8 connected paralleland having its impedance value Z@ and its resistance value Re.

The detector plate current flowing through plate circuit impedance Zp,which may be a resistor or other suitable impedance of impedance valueZp and resistance value Rp, sets up a voltage between amplifier inputterminals II and I2 which, after amplicatio-n, appears across terminalsI3 and I4 and thus across cathode impedance Ze. The sense of thisamplified voltage is such as to drive tube 2 toward cut-off and thusonly a small plate current is required to set up the 11o-signal bias.Let impedance Z0' be defined as the ratio Ec/Ic, where E@ is the biasdeveloped across ZC and Ic is the detector tube plate-cathode current.Then this apparent impedance will be greater than the actual value of Zcand its magnitude will be a measure of the effectiveness of the feedbackin reducing the necessary detector current. The magnitude of Ze may becalculated in terms of Zp, Ze, A, i and an additional impedance Z0.Impedance Z is the impedance measured between terminals I3 and I4, withthe connections to Ze broken at XX. During this measurement the outputload I must be left connected. The apparent impedance Ze' is then:

where Ai is the voltage gain of the ampliiierattenuator combinationmeasured between input terminals Il and I2 and output terminals I3 andI4, with the load connected and with the leads to Ze broken at XX. Thefactor in parentheses evidently multiplies the Value of Ze up to itsapparent value Ze. This expression is evidently valid at any frequencyand reduces to the form at zero frequency.

Having established the value of apparent im-` pedance in the cathodelead of the detector tube, the analysis of the system becomes relativelysimple. When a steady carrier is applied through input circuit I anadditional direct current component of current flows through tube 2, ofsufcient magnitude to develop across Ze a voltage E very nearly equal tothe radio frequency voltage between grid and cathode of tube 2. (Ze isarranged to include a path, as for example the path through condenser 8,of low impedance at the carrier frequency so that practically the entireapplied voltage appears between the grid and cathode of the tube.) Thusthere are now two components of current through the detector tube, theno-signal component and the component due to the signal carrier. Thesetwo components may be written as E lod-F where Eb is the volta-gedeveloped across Ze, or

between grid and cathode of tube 2', in the aby sence of signal.

When the carrier is modulated sinusoidally with a modulation factor 7c,the voltage across Ze must vary proportionally and a third component ofcurrent therefore must flow in tube 2. This component is of envelopefrequency and has an amplitude comparing this criterion with that forthe diode, it will be observed that the existence of the factor E/Eb-l-Etends to reduce the diiculty of preventing non-tracking distortion.

The circuit shown could be modified in several ways without changing itsmethod of operation. For example, attenuator i might be omitted, or theoutput obtained from terminals I3 and I4 instead of as shown.

More important modications would be required in order to make theamplifier responsive only to alternating voltages. In that case,blocking condensers (not shown) could be inserted at the points markedII and I3. Then the direct current path through Ze should be made tohave a high resistance Re, comparable in magnitude to Zo', as computedfor envelope frequencies in the required range. This computation shouldbe made from the same formulas as for the direct current amplifier. Thenon-tracking criterion for this case is rn. E Z E11-l E- From thisexpression, it will be evident that Re should be about equal to lZc'l.It should not exceed IZcI by more than, say, a factor of 2.

Radio frequency chokes IS and I1 prevent the incoming radio frequencywaves from bein-g fed back or from reaching the amplifier A.

In Fig. 5A, the amplifier is effective in reducing the current throughtube 2 in much the same way as in Fig. 5. The most important differencebetween the two circuits is that the envelope frequency bias voltage setup by the signal appears on the grid in Fig. 5A and on the cathode inFig. 5. -Since Fig. 5A is drawnl for the case in which the amplifier isisolated from direct voltages, the direct current biases due to therio-signal and carrier components of current are developed on thecathode, across resistance R2.

Blocking condensers CB isolate the amplifier from direct currents, andcondenser C2 by-passes resistance R2. These three condensers all havelow reactanc-e for envelope frequencies and `C2 must also have a lowreactance at radio frequencies. Condenser CR must have a high reactancefor envelope frequencies and a low reactance for radio frequencies. Theresistance Rg is -a high resistance grid leak used to maintain the -gridof tube 2 at a definite average voltage C-. .The zero-signal current maybe adjusted by Varying the Voltage C-. Since resistance R2 is large, thenecessary zero signal current tube 2' is small.

When a steady carrier is applied to tube 2 through tuned circuit I, thecurrent through R2 increases an amount sufficient to give an increase involtage across R2 nearly equal to the magnitude E of thecarrier voltage.The two direct current components of detector current are therefore E Ereife where Eb is the zero signal bias across R2.

The envelope frequency component of detector current must be of suchmagnitude as to develop a voltage of magnitude almost equal to lcE onthe grid. This requires a current Ip through Zp and tube 2 of magnitudeFrom this expression, it is evident that an apparent transfer impedanceZ may be defined such that Z=l.cE/Ip where lsE is the magnitude of theinput modulation envelope and Ip is the envelope frequency component ofdetector current. In that case Z'=ZpA1 and the criterion for nonon-tracking is sistance R. With these definitions made, the criterionfor no non-tracking becomes @gil [Z Eb-i-E The circuit of Fig. 5Bdiffers from both the preceding circuits in that the amplifier inputvoltage is obtained from an impedance in the cathode lead, rather thanthe plate lead, of tube 2'. Otherwise, it most nearly resembles Fig. 5A.However, the amplifier in Fig. 5B must have essentially degree phaseshift throughout the envelope frequency band, as opposed to Zero degreesin Fig. 5A.

In the form shown, the circuit is adapted for a direct currentamplifier. In that case, the zero-signal current through tube 2 dependson the characteristics of the tube 2 and of amplifier A and attenuatori. However, the entire detector tube current flows through impedance Zcand the bias voltage Eb which will appear in the no non-trackingcriterion is the voltage appearing across Ze in the absence of signal.

The apparent impedance Ze' which limits the fiow of detector tubeplate-cathode current is 'Ilhis reduces to Re' at zero frequency.Following the same type of reasoning as outlined above, the criterionfor no non-tracking is where al1 the symbols have already been defined.r

The circuit of Fig. 5B may be converted for the use of an amplifierisolated from direct voltages by following the general procedure alreadydescribed. Blocking condensers, as shown at CB in Fig. 5A should beinserted at the points marked B in Fig. 5B and a grid leak resistor andC-supply arranged to hold the grid of tube 2 in Fig. 5B-at Vthe desiredsteady bias Voltage as resistor Rg and the C-supply in Fig. 5A bias thegrid of tube 2' in Fig. 5A. The Vdirect current path in Ze should bemade a high resistance of magnitude approximately equal to Ze. Apparentimpedance Ze should be calculated in the same way as before, and thecriterion for no non-tracking becomes i cRc E' WIEN-E51 where Re is theresistance of the direct current path in Ze.

Fig. 5C shows a form of cathode-lead network suitable as la substitutefor that shown between point P andY ground in Fig. 5B when the amplieris to be an alternating current amplifier t(i. e., when the amplifier isto be isolated for direct current). In Fig. 5C the cathode leadimpedance comprises resistances Y 9 and Ill in series, with aradio-frequency by-pass condenser I8 `(of high reactance at the envelopefrequencies) connected across both 9 and l0, and with The magnitude ofthe age Eo is given by Ec=(Is-I1)Z (2) where I1 is the carrier frequencycomponent of the diode current with switch 22 closed and switch 2| open.Thus,

and under these conditions the average value of diode current will becarrier frequency voltwhere En is the voltage of bias source 40 and R isthe resistance of resistance 28, the factor MA having already beendefined. This relation expresses the fact that there are two directcurrent components of diode current, the rst being that sufficient tobuild up a negative bias on the diode approximately equal to themagnitude of voltage Ee, while the second is the component of currentdue to the bias source 40. From these four equations, inserting therequirement that Eo must be at least twice Ee, the following criterionfor the adjustment of resistance R and impedance Z may be established: i

MR l-ZTE l (5) Observance of this criterion will insure that properoperating conditions are obtained for the case of an unmodulatedcarrier. In order to determine the conditions necessary for operationwith switch 2| closed and a sinusoidal modulating voltage supplied atthe terminals Vof the bias source, it will be necessary to consider themagnitude of the envelope frequency component of current through thediode. Using double primes to represent quantities evaluated at envelopeor side band frequencies, this current becomes 121%? 4 6) where lc isthe modulation factor for the resultant modulated frequency voltageappearing across the diode. The condition necessary for no nontrackingdistortion may therefore be written 5- IZ//I EblZ//l 1;;- MkR nMkR l (7)which reduces to MICR E@ f wrm-1 y (8)..

I-t will be observed that this criterion is very similar to thosedeveloped for the preceding circuits. The factor M will bleapproximately one-half and its actual value may be determined bymeasurement of the fundamental component of current to the diodecompared with the resultant average current through resistor 2|3.l Inmaking this measurement, it will be advisable to reduce the voltage Etfrom the bias source 40 to zero.

If `switch 22 is open while the system lis operating in accordance withthe above description, the signal frequency voltage appearing acrossresistor 23 may be used to monitor the operation of the system. Resistor23 should then have a magnitude as small as is consistent with thedesired sensitivity of the monitor.

With switch 2| open, and switch 24 operated to replace source 20 by thedouble sideband modulation generator 29 or source of double sidebandmodulated signal, the system may be operated as a detector. In that casea condenser M3 having a low reactance at signal frequencies should beconnected in parallel with resistance 28 as by switch 47. A steady biaswill then appear across this condenser and resistance 28, approximatelyequal to the carrier value of the voltage appearing across the diode.When this carrier is modulated at signal frequency, an additionalcurrent, varying in magnitude at signal frequency, will flow through thediode and impedance 32. This current, consisting of a series of pulsesat carrier frequency, will tend to hold the diode radiofrequency voltageconstant through the combined action of impedance 32, amplifier A andattenuator ,6. Thus the current through the diode will contain acomponent whose magnitude is proportional to the modulation on the inputsignal. If switch 22 is opened, a voltage corresponding to the signalwill therefore be developed across resistance 23.

Under these conditions the device will also op'- erate as a limiter. Inother words, the radiofrequency voltage delivered at the output of theradio-frequency amplifier will be held nearly constant, despitevariations in the voltage delivered from the source.

Fig. 7 shows a detector circuit 50, generally similar to that of Fig. 3,in an envelope frequency negative feedback system comprising radiotransmitter 5| (including carrier oscillation generating and modulatingmeans) and signal input amplifier 52 for amplifying signals supplied toits input transformer 53 by signal input circuit 54 and transmitting theamplified signals through connection 5l to the modulator in the radiotransmitter to modulate the carrier oscillations generated in the radiotransmitter. The modulated carrier wave generated by the combination ofthe carrier oscillations and the signals in the modulator is transmittedfrom the radio transmitter to a radio-frequency load circuit 55, forexample, an antenna circuit.

The detector circuit 5i] is shown with its input connected across thecircuit 55 and its output connected across resistor 56 which is inseries with-the secondary winding of the signal input transformer 53 inthe input circuit of the signal amplifier 52. Modulated waves, from themodulator in the transmitter 5|, are demodulated in detector 50 and theresulting signal waves are fed back through connection 58 to the inputof amplifier'V 52 in phase opposition to signal waves supplied to theamplier from circuit 54. Thus, there is negative feedback around thefeedback loop including the modulator in the radio transmitter 5|, thedetector 50, and the amplier 52. In other words there is feedback aroundthis loop in the proper phase and amplitude to obtain the improvementsin distortion, noise, etc., which accrue from the application ofnegative feedback.

The amount of this negative feedback around this envelope feedback loopmay be large, as for example several times 10 decibels. However, inAenvelope-,frequency feedback operation, where a frequency change occursat the modulator in the Inu-circuit of the feedback loop, a limitationon the amount of improvement obtainable (in envelope distortion, etc.)by large amounts of feedback around the loop is encountered due to thepresence of a detector or demodulator; because inasmuch as thedemodulator is in the beta-circuit of this feedback loop, any distortiongenerated in the demodulator is present in the transmitter output. Thusthe problem of reducing non-linear distortion at the transmitter is notmerely a. matter of obtaining sufficient envelope feedback around thisloop, but involves obtaining a demodulator whose distortion issufficiently low to realize the improvement of transmitter distortionpossible with the increased amount of feedback. Moreover, if a usualtype of high level linear rectifier were used, it would need to have alarge filament emission with small transmit time and Work into a highimpedance load, and these requirements would not be compatible withthose for the maximum amount of feedback, which are low tube capacitanceand low output impedance. However, all these requirements are satisfiedby the detector t in the system of Fig. 7. This detector is a circuitwhich includes a linear rectifier, the diode I, and effectively applieslocal negative feedback to the diode through linear amplifier A in thegeneral fashion described above in connection With Fig. 3, for example.This circuit allows the use of a small rectifier tube with small transittime, low capacitance and only moderate lament emission working into alow output impedance. This feedback makes the rectifier operate as if ithad a high output impedance (load impedance) so that large filamentemission i's not required for small distortion. Though Without thisfeedback a very high physical resistance might be used, it would soincrease the time constant that non-tracking in the diode circuit mightresult. The feedback in the detector may be, for example, several times10 decibels, and reduces the time constant by a large factor. Thus thefidelity of the detector is improved by local feedback in thedemodulator which so modifies the impedance presented to the diode as tomaintain its fidelity even when the transmitter is deeply modulated athigh envelope frequencies.

In Fig. '7, condensers CB are blocking condensers. Co is a blockingcondenser for envelope frequencies. Circuit l comprising parallelconnected inductance, capacity and resistance is tuned to the radiocarrier frequency, fo, which may be for example of the order of 162megacycles, the signal being for example a group of 12 carrier telephonemessages occupying the frequency band extending from 60 kilocycles to108 kilocycles. This circuit i acts as a shunt to the signal frequencythat is present along with the modulated radio Wave on the modulatorplate. It helps to insure that the output will be fed back by means ofmodulation and demodulation and not directly at the signal frequency,its inductance presenting a very low impedance at the envelope frequencyso as to minimize any distortion pick-up of envelope voltage from themodulating amplifier. Also, it increases the input impedance of thedetector by anti-resonating the tube and the stray capacities.

R1 and C1 are the resistance and condenser across which the envelopevoltage is developed that is to be amplified by A'. The phase of theenvelope voltage developed in the plate circuit of A is such as toproduce the desired local feedback for the diode, the plate of A beingcoupled to the plate of the diode by connection which includes ablocking condenser CB and a, circuit 3 tuned to fu. The circuit 3isolates the plate of the diode from the envelope frequency amplifier atradio frequencies. It prevents the radio frequency from getting on theplate of amplifier A and helps prevent it from getting through to thegrid of the first tube of the signal-frequency amplifier 52.Radio-frequency choke 6I and series-tuned circuit 62 which is tuned tofo serve as a filter for the grid of amplifier A', minimizing theradio-frequency carrier voltage which otherwise might appear on the gridof A and cause overloading. Radio-frequency choke 10 further reduces theradio-frequency voltage on the grid of A. Choke B3, anti-resonant to fo,further suppresses the radio-frequency voltage to a value which will notcause distortion on the first signal-frequency amplifier grid. ResistorB4 is a grid bias and local negative feedback resistor for A. Condenser65 cooperates With resistance Rp to filter the plate current for tubeA'. Resistances 66 and 61 form a potential -divider for the screen gridcircuit of the tube.

Elements 65, 65 and 68 filter the screen grid current. Resistance R2corresponds to the resistance R2 in Fig. 3. Network 12 is part of theequalizing network needed in the main feedback loop. It has nosignificant effect on the operation of the demodulator circuit.

In the case of the detector 5i) as well as others of the detectorsdescribed above, an advantageous feature is that it is not necessarythat the radio-frequency feeding circuit have low impedance for themodulation frequencies.

What is claimed is:

1. A wave translating system comprising a source of signal modulatedcarrier Waves, a detector for demodulating said waves, an amplifierresponsive to signal energy from said detector, means for transmittingsignal energy from said detector to said amplifier, means connectingsaid source across a portion of the output circuit of said amplifier,and means for reducing the signal current in said detector comprisingmeans for transmitting signal energy from the output circuit of saidamplifier to said detector.

2. A demodulating system comprising a source of signal modulated carrierwaves, a detectoramplifier circuit comprising a detector and a signalamplifier, said detector-amplifier circuit having an input circuit andan amplifier output circuit, means for connecting said source acrosssaid input circuit, and means for applying signal waves from a portionof said output circuit across said input circuit, the phase shiftthrough said amplifier having such value as to produce negative feedbackof signal waves from said output circuit across said input circuit.

3. A demodulating system comprising a source of signal modulated carrierwaves, a detector, a signal amplifier having an input circuit and anoutput circuit, means for connecting said source across said detectorand said input circuit in series, and signal transmitting means forconnecting a portion of said output circuit across said detector andsaid input circuit in series, the phase shift through said amplierhaving such value as to produce negative feedback of signal waves fromits output circuit to its input circuit through said detector.

4. A demodulating system comprising a source of signal modulated carrierwaves, a detector, a detector load circuit including an amplifier forsignal Waves having an input circuit and an output circuit, a circuitfor transmitting signal Waves including said detector, said inputcircuit and a portion of said output circuit in series, and meansconnecting said source across said portion, said amplifier having itsphase shift such that it produces negative feedback of signal Waves fromits output circuit to its input circuit through said detector andnegative biasing voltage for the detector varying in accordance with thesignal and tending to reduce signal current flow through the detectorresulting from detection of modulated Waves supplied to said detectorfrom said source.

5. A Wave translating system comprising a source of Waves, a rectifier,an amplifier, means transmitting energy from said rectifier to saidamplifier, negative feedback means for said amplifier connecting inserial relation the amplier input circuit, a path through said rectifierand a path forming a portion of the amplifier output circuit, and meansconnecting said source across one of said paths.

6. A demodulating circuit comprising a source of signal modulatedcarrier waves, a diode, an amplifier responsive to signal Waves, andmeans for feeding said diode from said source and said amplier from saiddiode and producing negative feedback of signal Waves in said amplifier,comprising means connecting in serial relation said diode, the inputcircuit of said amplifier and a branched circuit including in one branchsaid source and in another branch a portion of the output circuit ofsaid amplifier.

7. A Wave translating system comprising a source of signal modulatedcarrier waves to be demodulated, a diode detector, an amplifier, meansfor transmitting signal Waves from said detector to the input circuit ofsaid amplifier, negative feedback means for said amplifier connecting inserial relation the input circuit of said amplifier, a path through saidrectifier and a portion of the output circuit of -said amplifier, andmeans connecting said source across said portion.

8. A Wave translating system comprising a source of signal modulatedcarrier Waves, a detector comprising an anode, a cathode and a controlelectrode connected to said source for demodulating said Waves, anamplifier responsive to signal energy from said detector, meanstransmitting signal energy from said detector to said amplier, and meansfor producing negative feedback of signal energy from the output circuitof said amplifier to said control electrode and cathode of saiddetector.

JAMES W. MCRAE.

